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mic2171 100khz 2.5a switching regulator micrel inc. ? 2180 fortune drive ? san jose, ca 95131 ? usa ? tel +1 ( 408 ) 944-0800 ? fax + 1 (408) 474-1000 ? http://www.micrel.com may 2007 1 m9999-051107 general description the mic2171 is a complete 100khz smps current-mode controller with an internal 65v 2.5a power switch. although primarily intended for voltage step-up applica- tions, the floating switch architecture of the mic2171 makes it practical for step-dow n, inverting, and cuk config- urations as well as isolated topologies. operating from 3v to 40v, the mic2171 draws only 7ma of quiescent current, making it attractive for battery operated supplies. the mic2171 is available in a 5-pin to-220 or to-263 for ?40c to +85c operation. data sheets and support doc umentation can be found on micrel?s web site at: www.micrel.com. features ? 2.5a, 65v internal switch rating ? 3v to 40v input voltage range ? current-mode operation, 2.5a peak ? internal cycle-by-cycle current limit ? thermal shutdown ? twice the frequency of the lm2577 ? low external parts count ? operates in most switching topologies ? 7ma quiescent current (operating) ? fits lt1171/lm2577 to-220 and to-263 sockets applications ? laptop/palmtop computers ? battery operated equipment ? hand-held instruments ? off-line converter up to 50w(requires external power switch) ? pre-driver for hi gher power capability ___________________________________________________________________________________________________________ typical application +5v (4.75v min.) d1 1n5822 mic2171 in sw fb c1* 47f r3 1k c3 1f r1 10.7k 1% c2 470f v out +12v, 0.25 a l1 15h * locate near mic2171 when supply leads > 2? r2 1.24k 1% gnd comp c4 470f d2 1n5818 c1 47f c2 1f r3 1k r4* v out 5v, 0.5 a * optional voltage clipper (may be req?d if t1 leakage inductance too high) c3* d1* r1 3.74k 1% r2 1.24k 1% v in 4v to 6v t1 1:1.25 l pri = 12h mic2171 in sw fb gnd comp figure 1. mic2171 5v to 12v boost converter figure 2. mic2171 5vflyback converter
micrel, inc. mic2171 may 2007 2 m9999-051107 ordering information part number standard rohs compliant* temperature range package mic2171bt mic2171wt ?40 to +85c 5-pin to-220 mic2171bu MIC2171WU ?40 to +85c 5-pin to-263 *rohs compliant with "high-melting solder" exemption. pin configuration tab gnd 5in 4sw 3gnd 2fb 1comp tab gnd 5 i n 4sw 3gnd 2fb 1comp 5-pin to-220 (t) 5-pin to-263 (u) pin description pin number pin name pin function 1 comp frequency compensation: output of tr ansconductance-type error amplifier. primary function is for loop stabilization. can also be used for output voltage soft-start and current limit tailoring. 2 fb feedback: inverting input of error amplifie r. connect to external resistive divider to set power supply output voltage. 3 gnd ground: connect directly to the input fi lter capacitor for proper operation (see applications info). 4 sw power switch collector: collector of np n switch. connect to external inductor or input voltage depending on circuit topology. 5 in supply voltage: 3.0v to 40v
micrel, inc. mic2171 may 2007 3 m9999-051107 absolute maximum ratings supply voltage (v in ) .......................................................40v switch voltage (v sw )......................................................65v feedback voltage (transient, 1ms) (v fb ) .....................15v lead temperature (solde ring, 10 se c.)...................... 300c storage temperature (t s ) .........................?65c to +150c esd rating (1) operating ratings operating temperature range ................... ?40 c to +85c junction temperature (t j ) ........................ ?55c to +150c thermal resistance to-220-5 ( ja ) (2) ...............................................45c/w to-263-5 ( ja ) (3) ................................................45c/w electrical characteristics v in = 5v; t a = 25c, bold values indicate ?40c< t a < +85c, unless noted. parameter condition min typ max units reference section feedback voltage (v fb ) v comp = 1.24v 1.220 1.214 1.240 1.264 1.274 v v feedback voltage line regulation 3v v in 40v v comp = 1.24v 0.6 %/v feedback bias current (i fb ) v fb = 1.24v 310 750 1100 na na error amplifier section transconductance (g m ) ? i comp = 25a 3.0 2.4 3.9 6.0 7.0 a/mv a/mv voltage gain (a v ) 0.9v v comp 1.4v 400 800 2000 v/v output current v comp = 1.5v 125 100 175 350 400 a a output swing high clamp, v fb = 1v low clamp, v fb = 1.5v 1.8 0.25 2.1 0.35 2.3 0.52 v v compensation pin threshold duty cycle = 0 0.8 0.6 0.9 1.08 1.25 v v output switch section on resistance i sw = 2a, v fb = 0.8v 0.37 0.50 0.55 ? ? current limit duty cycle = 50%, t j 25c duty cycle = 50%, t j < 25c duty cycle = 80%, note 4 2.5 2.5 2.5 3.6 4.0 3.0 5.0 5.5 5.0 a a a breakdown voltage (bv) 3v v in 40v i sw = 5ma 65 75 v oscillator section frequency (f o ) 88 85 100 112 115 khz khz duty cycle [ (max)] 80 90 95 % input supply voltage section minimum operating voltage 2.7 3.0 v quiescent current (i q ) 3v v in 40v, v comp = 0.6v, i sw = 0 7 9 ma supply current increase ( ? i in ) ? i sw = 2a, v comp = 1.5v, during switch on-time 9 20 ma notes: 1. devices are esd sensitive. handling precautions recommended. 2. mounted vertically, no external heat sink, 1/4 inch leads soldered to pc board containing approximately 4 inch squared copp er area surrounding leads. 3. all ground leads soldered to approximately 2 in ches squared of horizontal pc board copper area. 4. for duty cycles ( ) between 50% and 95%, minimum guaranteed switch current is i cl = 1.66 (2- ) amp (pk).
micrel, inc. mic2171 may 2007 4 m9999-051107 typical characteristics 2.3 2.4 2.5 2.6 2.7 2.8 2.9 -100 -50 0 50 100 150 minimum operating voltage (v) temperature (c) minimum operating voltage switch current = 2a 0 100 200 300 400 500 600 700 800 -100 -50 0 50 100 150 feedback bias current (na) temperature (c) feedback bias current -5 -4 -3 -2 -1 0 1 2 3 4 5 0 10203040 feedback voltage change (mv) v in operating (v) feedback voltage line regulation t j =-40c t j =25c t j = 125c 5 6 7 8 9 10 11 12 13 14 15 0 10203040 supply current (ma) v in operating voltage (v) supply current i sw =0 d.c.= 90% d.c.= 50% d.c.= 0% 0 10 20 30 40 50 01234 average supply current (ma) switch current (a) supply current = 90% = 50% 0 1 2 3 4 5 6 7 8 9 10 -100 -50 0 50 100 150 supply current (ma) temperature(c) supply current v comp =0.6v 0 0.2 0.4 0.6 0.8 1.0 1.2 1.4 1.6 0123 switch on voltage (v) switch current (a) t j =25c t j =125c t j =?40c switch on-voltage 60 70 80 90 100 110 120 -50 0 50 100 150 frequency (khz) temperature(c) oscillator frequency 0 2 4 6 8 0 20406080100 switch current (a) duty cycle (%) current limit ?40c 25c 125c error amplifier gain 0.0 0.5 1.0 1.5 2.0 2.5 3.0 3.5 4.0 4.5 5.0 -100 -50 0 50 100 150 transconductance (a/mv) temperature(c) 0 1000 2000 3000 4000 5000 6000 7000 1 10 100 1000 10000 transconductance (s) frequency (khz) error amplifier gain 210 180 150 120 90 60 30 0 -30 1 10 100 1000 10000 phase shift () frequency (khz) error amplifier phase
micrel, inc. mic2171 may 2007 5 m9999-051107 functional diagram current amp. error amp. 1.24v ref. logic driver anti-sat. sw 100khz osc. reg. fb comp gnd in d1 q1 2.3v com- parator functional description refer to ?block diagram mic2171?. internal power the mic2171 operates when v in is 2.6v. an internal 2.3v regulator supplies biasing to all internal circuitry including a precision 1.24v band gap reference. pwm operation the 100khz oscillator generat es a signal with a duty cycle of approximatel y 90%. the current-mode comparator output is used to reduce the duty cycle when the current amplifier output voltage exceeds the error amplifier output voltage. the resulting pwm signal controls a driver which supplies base current to output transistor q1. current-mode advantages the mic2171 operates in current mode rather than voltage mode. there are three distinct advantages to this technique. feedback loop compensation is greatly simplified because inductor current sensing removes a pole from the closed loop response. inherent cycle-by- cycle current limiting greatly improves the power switch reliability and provides automat ic output current limiting. finally, current-mode operation provides automatic input voltage feed forward which pr events instantaneous input voltage changes from disturbing the output voltage setting. anti-saturation the anti-saturation diode (d1) increases the usable duty cycle range of the mic2171 by eliminating the base to collector stored charge which would delay q1?s turnoff. compensation loop stability compensation of the mic2171 can be accomplished by connecting an appropriate network from either comp to circuit ground (see ?typical applications?) or comp to fb. the error amplifier output (comp) is also useful for soft start and current limiting. because the error amplifier output is a transconductance type, the output impedance is relatively high which means the output voltage can be easily clamped or adjusted externally.
micrel, inc. mic2171 may 2007 6 m9999-051107 application information soft start a diode-coupled capacitor from comp to circuit ground slows the output voltage rise at turn on (figure 3). mic2171 in comp c2 r1 v in d2 c1 d1 figure 3. soft start the additional time it takes for the error amplifier to charge the capacitor corresponds to the time it takes the output to reach regulation. diode d1 discharges c1 when v in is removed. current limit mic2171 in comp c2 r3 v in q1 r2 gnd r1 c1 sw fb v out note: input and output returns not commo n i cl 0.6v/r2 figure 4. current limit the maximum current limit of the mic2171 can be reduced by adding a voltage clamp to the comp output (figure 4). this feature c an be useful in applications requiring either a complete shutdown of q1?s switching action or a form of current fold-back limiting. this use of the comp output does not disable the oscillator, amplifiers or other circuitry, therefore, the supply current is never less than approximately 5ma. thermal management although the mic2171 family contains thermal protection circuitry, for best reliabilit y, avoid prol onged operation with junction temperatures near the rated maximum. the junction temperature is determined by first calculating the power dissipat ion of the device. for the mic2171, the total power dissipation is the sum of the device operating losses and power switch losses. the device operating losses are the dc losses associated with biasing all of the internal functions plus the losses of the power switch driver circuitry. the dc losses are calculated from the supply voltage (v in ) and device supply current (i q ).the mic2171 supply current is almost constant regardless of the supply voltage (see ?electrical characteristics?). the driver section losses (not including the switch) are a function of supply voltage, power switch current, and duty cycle. p (bias+driver) = (v in i q ) + (v in(min) x i sw x ? i in ) where: p (bias+driver) = device operating losses v in(min) = supply voltage = v in ? v sw i q = typical quiescent supply current i cl = power switch current limit ? i in = typical supply current increase as a practical example refer to figure 1. v in = 5.0v i q = 0.007a i cl = 2.21a = 66.2% (0.662) then: v in(min) = 5.0v ? (2.21 x 0.37) = 4.18v p (bias+driver) = (5 x 0.007) + (4.18 x 2.21 x 0.009) p (bias+driver) = 0.1w power switch dissipation calculations are greatly simplified by making two assumptions which are usually fairly accurate. first, the majority of losses in the power switch are due to on-losses. to find these losses, assign a resistance value to the collector/emitter terminals of the device using the saturation voltage versus collector current curves (see typical performance character- istics). power switch losses are calculated by modeling the switch as a resistor with the switch duty cycle modifying the average power dissipation. p sw = (i sw ) 2 r sw where: = duty cycle f out in(min) f out v v v v v + ? + = v sw = i cl (r sw ) v out = output voltage v f = d1 forward voltage drop at i out from the typical perfor mance characteristics: r sw = 0.37 ? then: p sw = (2.21) 2 0.37 0.662 p sw = 1.2w p (total) = 1.2 + 0.1 p (total) = 1.3w
micrel, inc. mic2171 may 2007 7 m9999-051107 the junction temperature for any semiconductor is calculated using the following: t j = t a + p (total) ja where: t j = junction temperature t a = ambient temperature (maximum) p (total) = total power dissipation ja = junction to ambient thermal resistance for the practical example: t a = 70c ja = 45c/w (to-220) then: t j = 70 + (1.24 45) t j = 126c this junction temperature is below the rated maximum of 150c. grounding refer to figure 5. heavy lines indicate high current paths. mic2171 in sw fb vc v in gnd single point ground figure 5. single point ground a single point ground is strongly recommended for proper operation. the signal ground, compensation network ground, and feed-back network connections are sensitive to minor voltage variations. the i nput and output capacitor grounds and power ground conductors will exhibit voltage drop when carrying large currents. keep the sensitive circuit ground traces separate from the power ground traces. small voltage variations applied to the sensitive circuits can pr event the mic2171 or any switching regulator from functioning properly. boost conversion refer to figure 1 for a typical boost conversion application where a +5v logic supply is available but +12v at 0.25a is required. the first step in designing a boost converter is determining whether inductor l1 will cause the converter to operate in either contin uous or discontinuous mode. discontinuous mode is preferred because the feedback control of the converter is simpler. when l1 discharges its current completely during the mic2171 off-time, it is operating in discontinuous mode. l1 is operating in conti nuous mode if it does not discharge completely before the mic2171 power switch is turned on again. discontinuous mode design given the maximum output current, solve equation (1) to determine whether the device can operate in discontinuous mode without initiating the internal device current limit. (1) out in(min) cl out v v 2 i i ? ? ? ? ? ? ? ? (1a) f out in(min) f out v v v v v + ? + = where: i cl = internal switch current limit i cl = 2.5a when < 50% i cl = 1.67 (2 ? ) when 50% (refer to electrical characteristics.) i out = maximum output current v in(min) = minimum input voltage = v in ? v sw = duty cycle v out = required output voltage v f = d1 forward voltage drop for the example in figure 1. i out = 0.25a i cl = 1.67 (2?0.662) = 2.24a v in(min) = 4.18v = 0.662 v out = 12.0v v f = 0.36v (@ .26a, 70c) then: 12 0.662 4.178 2 2.235 i out ? ? ? ? ? ? i out 0.258a this value is greater than the 0.25a output current requirement, so we can proceed to find the minimum inductance value of l1 for discontinuous operation at p out . (2) sw out 2 in f 2p ) (v l1
micrel, inc. mic2171 may 2007 8 m9999-051107 where: p out = 12 0.25 = 3w f sw = 1105hz (100khz) for our practical example: () 5 2 10 1 3.0 2 0.662 4.178 l1 l1 12.4h (use 15h) equation (3) solves for l1?s maximum current value. (3) l1 t v i on in l1(peak) = where: t on = / fsw = 6.6210-6 sec 6 6 l1(peak) 10 15 10 6.62 4.178 i ? ? = i l1(peak) = 1.84a use a 15h inductor with a peak current rating of at least 2a. flyback conversion flyback converter topology may be used in low power applications where voltage isolation is required or whenever the input voltage can be less than or greater than the output voltage. as with the step-up converter the inductor (transformer primary) current can be continuous or discontinuous. discontinuous operation is recommended. figure 2 shows a practical flyback converter design using the mic2171. switch operation during q1?s on time (q1 is the internal npn transistor? see block diagrams), energy is stored in t1?s primary inductance. during q1?s o ff time, stored energy is partially discharged into c4 (output filter capacitor). careful selection of a low esr capacitor for c4 may provide satisfactory output ripple voltage making additional filter stages unnecessary. c1 (input capacitor) may be reduced or eliminated if the mic2171 is located near a low impedance voltage source. output diode the output diode allows t1 to store energy in its primary inductance (d2 non-conducting) and release energy into c4 (d2 conducting). the low forward voltage drop of a schottky diode minimizes power loss in d2. frequency compensation a simple frequency compensation network consisting of r3 and c2 prevents output oscillations. high impedance output stages (transconductance type) in the mic2171 often permit simplified loop-stability solutions to be connected to circuit ground, although a more conventional technique of connecting the components from the error amplifier output to its inverting input is also possible. voltage clipper care must be taken to minimize t1?s leakage inductance, otherwise it may be necessary to incorporate the voltage clipper consisting of d1, r4, and c3 to avoid second breakdown (failure) of the mic2171?s internal power switch. discontinuous mode design when designing a discontinuou s flyback converter, first determine whether the device can safely handle the peak primary current demand placed on it by the output power. equation (8) finds the maximum duty cycle required for a given input voltage and output power. if the duty cycle is greater than 0.8, discontinuous operation cannot be used. (8) () vsw v i 2p in(min) cl out ? for a practical example let: (see figure 2) p out = 5.0v 0.5a = 2.5w v in = 4.0v to 6.0v i cl = 2.5a when < 50% 1.67 (2 ? ) when 50% then: v in(min) = v in ? (i cl r sw v in(min) = 4 ? 0.78v v in(min) = 3.22v 0.74 (74%), less than 0.8 so discontinuous is permitted. a few iterations of equation (8) may be required if the duty cycle is found to be greater than 50%. calculate the maximum transformer turns ratio a, or n pri /n sec , that will guarantee safe operation of the mic2171 power switch. (9) sec in(max) ce ce v v f v a ? where: a = transformer maximum turns ratio v ce = power switch collector to emitter maximum voltage f ce = safety derating factor (0.8 for most commercial and industrial applications) v in(max) = maximum input voltage v sec = transformer secondary voltage (v out + v f )
micrel, inc. mic2171 may 2007 9 m9999-051107 for the practical example: v ce = 65v max. for the mic2171 f ce = 0.8 v sec = 5.6v then: 5.6 6.0 0.8 65 a ? a 8.2 (n pri /n sec ) next, calculate the maximum primary inductance required to store the needed output energy with a power switch duty cycle of 55%. (10) out 2 on 2 in(min) sw pri p t v 0.5f l where: l pri = maximum primary inductance f sw = device switching frequency (100khz) v in(min) = minimum input voltage t on = power switch on time then: 2.5 )2 10 (7.4 (3.22) 10 1 0.5 l 6 2 5 pri ? l pri 11.4h use a 12h primary inductance to overcome circuit inefficiencies. to complete the design the inductance value of the secondary is found which w ill guarantee that the energy stored in the transformer during the power switch on time will be completed discharg ed into the output during the off-time. this is nece ssary when operating in discontinuous-mode. (11) out 2 off 2 sec sw sec p t v 0.5f l where: l sec = maximum secondary inductance t off = power switch off time then: 2.5 ) 10 (2.6 (5.41) 10 1 0.5 l 2 6 2 5 sec ? l sec 7.9h finally, recalculate the transformer turns ratio to insure that it is less than the value earlier found in equation (9). (12) sec pri l l a then: 20 . 1 7.9 11.4 a = this ratio is less than the ratio calculated in equation (9). when specifying the transformer it is necessary to know the primary peak current which must be withstood without saturating the transformer core. (13) pri on in(min) peak(pri) l t v i = so: pri 6 peak(pri) l 10 7.6 3.22 i ? = i peak(pri) = 2.1a now find the minimum reverse voltage requirement for the output rectifier. this re ctifier must have an average current rating greater than the maximum output current of 0.5a. (14) a f a) (v v v br out in(max) br + where: v br = output rectifier maximum peak reverse voltage rating a = transformer turns ratio (1.2) f br = reverse voltage safety derating factor (0.8) then: 1.2 0.8 1.2) (5.0 6.0 v br + v br 12.5v a 1n5817 will safely handle voltage and current require- ments in this example. forward converters micrel?s mic2171 can be used in several circuit configurations to generate an output voltage which is less than the input voltage (bu ck or step-down topology). figure 6 shows the mic2171 in a voltage step-down application. because of the inte rnal architecture of these devices, more external components are required to implement a step-down regulator than with other devices offered by micrel (refer to the lm257x or mic457x family of buck switchers). however, for step-down conversion requiring a transformer (forward), the mic2171 is a good choice. a 12v to 5v step-down converter using transformer isolation (forward) is show n in figure 6. unlike the isolated flyback converter which stores energy in the primary inductance during the controller?s on-time and releases it to the load during the off-time, the forward converter transfers energy to the output during the on-
micrel, inc. mic2171 may 2007 10 m9999-051107 time, using the off-time to re set the transformer core. in the application shown, the trans former core is reset by the tertiary winding dischargi ng t1?s peak magnetizing current through d2. for most forward converters the duty cycle is limited to 50%, allowing the transformer flux to reset with only two times the input voltage appearing across the power switch. although during normal operation this circuit?s duty cycle is well below 50%, the mic2172 has a maximum duty cycle capab ility of 90%. if 90% was required during operation (start-up and high load currents), a complete reset of the transformer during the off-time would require the voltage across the power switch to be ten times the input voltage. this would limit the input voltage to 6v or less for forward converter applications. to prevent core saturation, the application given here uses a duty cycle limiter consisting of q1, c4 and r3. whenever the mic2171 exceeds a duty cycle of 50%, t1?s reset winding current turns q1 on. this action reduces the duty cycle of the mic2171 until t1 is able to reset during each cycle. d3 1n5819 mic2171 in sw fb comp c1 22f c3 1f r2 1k v out 5v, 1 a r4 3.74k 1% r5 1.24k 1% c5 470f l1 100h v in 12v gnd t1 1:1:1 d4 1n5819 d2 1n5819 * voltage clipper ? duty cycle limiter d1* c2* q1 ? c4 ? r3 ? r1* figure 6. mic2171 forward converter
micrel, inc. mic2171 may 2007 11 m9999-051107 package information 5-pin to-220 (t) 5-pin to-263 (u)
micrel, inc. mic2171 may 2007 12 m9999-051107 micrel, inc. 2180 fortune drive san jose, ca 95131 usa tel +1 (408) 944-0800 fax +1 (408) 474-1000 web http://www.micrel.com the information furnished by micrel in this data sheet is belie ved to be accurate and reliable. however, no responsibility is a ssumed by micrel for its use. micrel reserves the right to change circuitry and specifications at any time without notification to the customer. micrel products are not designed or authori zed for use as components in life support app liances, devices or systems where malfu nction of a product can reasonably be expected to result in personal injury. life suppo rt devices or systems are devices or systems that (a) are in tended for surgical implant into the body or (b) support or sustain life, and whose failure to perform can be reasonably expected to result in a significan t injury to the user. a purchaser?s use or sale of micrel produc ts for use in life support app liances, devices or systems is a purchaser?s own risk and purchaser agrees to fully indemnify micrel for any damages resulting from such use or sale. ? 2005 micrel, incorporated.

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